Currently, 3rd generation (3G) cellular communication systems are being developed to further enhance the communication services provided to mobile phone users. The most widely adopted 3rd generation communication systems are based on Code Division Multiple Access (CDMA) technology. Carrier frequencies are used for both uplink transmissions, i.e. transmissions from a mobile wireless communication unit (often referred to as wireless subscriber communication unit or user equipment in 3rd generation systems) to the communication infrastructure via a wireless serving base station (often referred to as a Node-B in 3rd generation systems) and downlink transmissions, i.e. transmissions from the communication infrastructure to the mobile wireless communication unit via a wireless serving base station (e.g. Node-B). A further description of CDMA, and specifically of the Wideband CDMA (WCDMA) mode of Universal Mobile Telecommunication System (UMTS), can be found in ‘WCDMA for UMTS’, Harri Holma (editor), Antti Toskala (Editor), Wiley & Sons, 2001, ISBN 0471486876.
CDMA communication, as used in 3G mobile communications air interface technologies, is an ‘interference limited’ technology from a data throughput perspective. CDMA technology utilises orthogonal variable spreading factor (OVSF) codes combined with pseudo noise (Pn) codes to differentiate multiple UEs that are utilising the same spectrum at the same time for uplink access on the Uu radio interface. In order to maintain sufficient signal-to-interference ratio (SIR) protection for all UEs on accessing the Node-B, up-link (UL) power control (PC) is dynamically managed by the network infrastructure. SIR estimation is commonly derived from pilot tones in the uplink (UL) dedicated paging control channel (DPCCH). User equipment (UE) devices transmitting to a Node-B on the same spreading factor (SF) would be arranged such that their respective transmissions have substantially the same power when received at the receiving Node-B. Often, up to ninety six UEs are simultaneously supported in call mode for a specific Node-B.
Modern modulation schemes used in many cellular communication systems use high peak-to-average ratios. A peak-to-average ratio of 10.5 dB is not uncommon in many versions of 3rd generation partnership project (3GPP) wireless communication systems, such as: EDGE, wideband code division multiple access (WCDMA), WiMAX and long term evolution (LTE). Therefore, the PA needs to be operating in a linear mode when using these modulation schemes, thereby driving down the PA efficiency to sub 10%. This implies that a 100 W PA consumes in excess of 1 kW DC power.
Major efforts have been underway in recent years to improve this poor power efficiency by utilising schemes such as adaptive predistortion. Predistortion schemes utilise feedback paths where the PA output is monitored and the resultant modulation signal and the distortion detection enables an ‘anti-distortion’ co-efficient to be applied to the (forward path) modulation signal, thereby compensating for (off-setting) the subsequent signal distortion created by the PA. In this manner, the use of predistortion schemes allows the PA to operate in a more non-linear mode of operation, thereby increasing the PAs overall efficiency. Thus, as a result of this efficiency drive, the selection and operation of the PA is closely coupled to the operation of the modulator components.
Conventional antenna arrays, comprising multiple antenna elements and used with existing Node-B equipment in most 3G installations, utilise a fixed +/−65° beam pattern. Outside of the main lobe of the antenna beam the signals are spatially filtered and significantly attenuated. Conventional network planning and passive antenna array solutions process all incoming signals with a common fixed beam pattern. Such receive processing, based on signals received within the geographic area identified by the antenna beam main lobe, referred to as the RF footprint, tends to dictate a corresponding common beam pattern for transmitter operation. Thus, an identical radio frequency (RF) footprint is used for both receive (Rx) and transmit (Tx) operation.
Rx beam-forming using antenna arrays depends on the ability to constructively add incident signals on each of the antenna elements in a way that coherently adds those from the desired direction. Thus, incident signals that are not from the desired direction will be incoherently added, and thus will not experience the same processing gain. The term ‘coherency’ implies that the signals will have substantially the same phase angle. In addition, thermal noise from multiple sources also exhibits incoherent properties, and thus when added the signals from multiple sources do not experience the same processing gain as a coherent desired signal.
Conversely in Tx active antenna arrays the signals are coherently combined within the intended beam pattern as electromagnetic (EM) signals in the ‘air’ so that they arrive coherently at the mobile station (MS) (e.g. UE) receiver.
In a Node-B antenna array arrangement, the received RF signal from a single UE cannot be discerned without demodulation of the composite signal. Individual receive beam-forming for a specific user is not feasible, since there is likely to be multiple received signals of comparable powers from different UEs simultaneously at the antenna array. Even if few UEs are utilising the Node-B, the likelihood is that the signals would be below the noise floor of the Node-B's receiver. The processing gain of a WCDMA receiver implies that the signal can be extracted from the noise floor. This, however, requires at least a partial demodulation process.
Most known beam-forming schemes consider beam-forming only in the radio frequency (RF) domain. Therefore, common baseband filter stages are used. The inventors of the present invention have recognised and appreciated that variation in the common baseband filter stages may dominate overall latency in the system, which have not been considered or corrected for in the known prior art. Furthermore, digital-to-analogue converters and analogue-to-digital converters are required to sample at increased frequency rates, driven by new air interface protocols. Thus, maintaining clock phase synchronization to such devices across the array of elements is becoming increasingly difficult. Furthermore, samples processed in the digital domain may be subject to latencies in excess of an integer cycle compounding the need to resolve latency mismatch.
For example, if a slow rate of change of modulation with respect to the RF is considered, then any latency in paths that correspond to multiple wavelengths of the RF signal have little effect. One example of this would be the case of the Global System for Mobile communications (GSM) standard, where the rate of change of phase per RF cycle is so low that it is not measured. In the GSM standard, the symbol rate is approximately 270 kS/s. In contrast, the more recent mobile communication technologies employ air-interface protocols that have symbol rates in wider bandwidth RF signals, where the rate of change of phase/amplitude per RF cycle is significant. Such a rate of change of phase/amplitude per RF cycle level has been found to cause distortion to the resultant beam-forming of signals.
WO 2008/000318(A1) highlights a problem of reference calibration signal generation and feedback path effects on phase/amplitude measurements. In order to attempt to solve this problem, WO 2008/000318(A1) proposes a coupler scheme that necessitates that multiple receiver or transmitter chains need to be disabled in order to perform the calibrations. This requirement is an artefact of the common coupler structure to multiple antenna paths. However, WO 2008/000318(A1) fails to describe a mechanism to perform calibration. It is noted that the proposed coupler scheme would also substantially degrade network performance when performed during live transmission of a network, as all but one transmitter at a time can be measured.
U.S. Pat. No. 6,339,399 B1 proposes a mechanism that uses distinct beam-forming component blocks to that for receive calibration resultant correction. In the mechanism proposed in U.S. Pat. No. 6,339,399 B1, only amplitude and phase correction on the respective receive paths is taken into consideration, due to the sole use of a complex multiplier as the corrective mechanism. In the mechanism proposed in U.S. Pat. No. 6,339,399 B1, only one coupling path per antenna element feed is employed. This, as highlighted in U.S. Pat. No. 6,339,399 B1, can cause an effect whereby feedback or calibration tone path error can dominate over receive path error.
Consequently, current techniques are suboptimal. Hence, an improved mechanism to address the problem of supporting antenna array technology in a wireless communication network would be advantageous.